Digital data over voice communication

ABSTRACT

A method and apparatus is described for transmitting and receiving data signals and voice band signals over a single pair of wires, wherein the energy content of the data signals in the voice band is transferred to a higher frequency to avoid interference between the two. This is accomplished by sinusoidally encoding the data pulses in the frequency domain. The encoding is equivalently performed in the time domain by linearly combining weighted delayed and advanced versions of the data pulses, in accordance with a weighting formula. A transversal filter is used to multiple delayed and advanced versions of the data pulses by a scaling factor times the ratio of m!/m-i)!i! factorial wherein i is the ith version being weighted, m is an integer greater than one and ! indicates the factorial function.

This application is a continuation of U.S. application Ser. No.07/159,887, filed Feb. 24, 1988.

TECHNICAL FIELD

This invention is in the field of digital data and voice communicationsystems over wire cable.

BACKGROUND ART

Telephone voice communication systems are ubiquitously deployed overmost of the United States and a good portion of the world. Telephonevoice communication uses the frequency band 300 Hz to 3300 Hz. A copperpair connects the end user's premises to the serving central office.This provides satisfactory voice signal transmission. As the need fordata communication increases, transmission of data signals from the enduser's premises to the serving central office, becomes a problem. Usingvoice band modems, the user can establish data communication up to about4.8 kbps, although 9.6 kbps modems are becoming available. These modemsrequire elaborate circuitry to condense an essentially wide band dataspectrum into the 300 Hz to 3300 Hz voice band. Also, when thecustomer's copper pair is being used for data communication, it is notavailable for voice communication. Not only does the narrow voice bandrestrict the rate at which data communication can occur, the use ofvoice band modems does not allow interactive voice/data transactions tooccur. Finally, since voice band data modems use the public switchednetwork; which was designed for voice communications, the degree ofperformance cannot be guaranteed.

Presently, high speed and high performance data applications requirethat the customer subscribe to a wide band four wire baseband (digital)data service, such as, Dataphone Digital Service (DDS). In DDS, highperformance is obtained by leaving the data un-modulated andtransmitting and receiving it through cable equalizers/pre-equalizers.In pending U.S. Pat. application Ser. No. 891,462 filed July 29, 1986,Gupta discloses that by use of a pre- and post-equalizer, the coppercable media can be effectively made wide band. Baseband data in the formof Pulse Amplitude Modulated Nyquist Pulses transmitted through thiscable equalizer combination results in good (≧70% open) eye performance.Thus, error rate performance better than 10⁻⁸ can be achieved. Athorough treatment of "eye opening" can be found in the book entitled"Data Transmission" by W.R. Bennett and J.R. Davey, page 119, McGrawHill, 1965.

The problems with this baseband Pulse Amplitude Modulation (PAM)technique are that (1) there is a D.C. component present and (2) thereare discrete frequencies present at the baud rate and its multiples.Since telephone copper cable is subject to lightning hits, powercrosses, etc., it is highly desirable to interface it throughtransformers and protection circuitry. The presence of D.C. in the PAMdata signal does not allow it to pass through a transformer. Also,energy concentrated at discrete frequency points, can cause cross-talkproblems in the other wire pairs in the binder group of which the PAMsignal carrying pairs are members. These two problems are solved by useof Alternate Mark Inversion (AMI) pre-coding, wherein the polarity ofevery alternate "1" is reversed to a "-1". Proof that this eliminatesD.C. and discrete frequencies is complex and can be found in "SignalTheory" by L.E. Franks, pages 217-218, Prentice Hall, 1969. Franksdemonstrates that the spectra of the AMI/PAM digital signal is of theform (sin Kf).×(f) which becomes zero, i.e., D.C. at f=0. In general,the slope at f=0 is not zero and, hence, significant energy is presentin the voice band (300 Hz to 3300 Hz). Thus, AMI/PAM and baseband voicetransmission are mutually incompatible.

With the explosion of data communication, a need exists for a method andapparatus for transmission and reception of voice and data signalsbetween terminals or nodes over a single pair of standard telephonecable. Present techniques for satisfying this need involve frequencyshift keying (FSK) in which two frequencies are required. The errorperformance of FSK transmission is unsatisfactory for a variety ofreasons. The high frequency band is highly attenuated by even moderatelength cable and so poor Signal to Noise Ratios (SNR) result leading todegraded performance. Furthermore, since the energy is clustered innarrow bands, cross-talk into other cables in the binder group iscreated. The more power used for transmission, the greater thecross-talk, limiting the use of other cables in the binder group forwideband services. Therefore, the signal cannot be transmitted as far asone would desire and administrative restrictions have to be placed onmixing other services in the same binder group. Expensive modulation anddemodulation circuits are also required for FSK and are a furtherdetriment.

DISCLOSURE OF THE INVENTION

In the apparatus of the present invention, a coding circuit is used toencode the data signal prior to transmission. The coding circuit encodesthe data signal in such a manner that the voiceband is vacated and thesignal energy is spread over a relatively broad frequency spectrum.Thus, the energy is not clustered in a narrow band and cross-talk isthereby minimized. Baseband transmission is employed so that signalerrors caused by FSK are avoided. No modulators or demodulators arerequired. The empty voiceband can then be used for baseband "Plain OldTelephone System" (POTS) communication.

The apparatus of the present invention accepts baseband data signalswhich are time compressed and multiplexed for burst or packetizedtransmission. Baseband POTS service can simultaneously be provided onthe same wire pair.

The time compressed and multiplexed pulse data signals are coupled to apre-coder which empties the energy content of the data signals from thevoiceband. This is accomplished by forming suitable weighted linearcombinations or summations of the pulses with delayed and advancedversions of the pulses. This process, as will be shown, is equivalent toencoding the Fourier transform of the pulses with a sinusoidal functionof the form sin^(m) θ; wherein θ=πf^(T/2), m is an integer greater thanor equal to 1, f is a frequency variable, and T is the reciprocal of thebaud rate. As "m" is increased, more and more pulse energy is removedfrom the low frequency band to a higher frequency band.

In a specific embodiment of the invention, m=4 resulting in sin⁴ θ,which equals a ##EQU1## pulse function shaping system. In a firstembodiment of the invention, the pre-coder comprises a rail former, atime domain filter and a summing circuit. The rail former divides theAMI/PAM coded baseband time-compressed multiplexed Nyquist or AlmostNyquist pulse data signals into two sets of data streams of sequentialalternate pulses from the input data stream. One data stream comprisespositive going pulses and the other comprises negative going pulses. Thedata streams are separately coupled to a time domain filter comprisingtwo shift registers with weighting resistors coupled to the outputstages. The shift registers provide delayed and advanced versions ofeach pulse. The resistors are chosen and coupled to a summing device toproduce a weighted sum voltage waveform having (1-cos2θ)² shaped pulsesin which the energy content of the pulses is spread over a frequencyrange higher than the frequency of the voiceband signals. If thebaseband signals are not AMI coded, only one data stream and one shiftregister is required for the time domain filter.

The shaped pulses are then coupled through a high pass filter, a lineimpedance matching resistor and a coupling transformer to the Tip andRing lines of a standard two wire telephone balanced line fortransmission to a substantially identical transmit/receive terminal atthe other end of the line. Voiceband signals are also coupled onto theTip and Ring lines through a passive low pass filter and transmittedover the same line. The received or incoming signals from the otherterminals are separated into low frequency (voiceband) and highfrequency (databand) signals by the a low pass filter and an additionalhigh pass filter in the receiver section. The data pulses are detectedand equalized for transmission losses over the cable and divided intopositive going and negative going received signals for demultiplexingand decoding. The invention will now be described in detail inconnection with the drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a plot of amplitude versus time for a Nyquist Pulse for thepurpose of illustrating principles of the invention.

FIG. 2 is an eye diagram, amplitude versus time plot of a "RaisedCosine" Nyquist pulse for illustrating features of the invention.

FIG. 3 is an illustration of a method for producing Almost NyquistPulses using a third order Butterworth Filter.

FIG. 4 is a plot of the power spectral density versus time of asinusoidally encoded Nyquist pulse with increasing order of the codingfunction "m".

FIG. 5 is a plot of amplitude versus time of a pulse waveform of thewave described by Equation 22₄.

FIG. 6 is a schematic diagram of a sinusoidal encoder in accordance withthe invention.

FIG. 7 is a schematic/logic diagram of a decoder for decodingsinusoidally encoded pulses S_(m) (t) wherein "m"=4.

FIG. 8 is a schematic of a concatenated encoding system wherein AMIencoding is followed by sinusoidal encoding.

FIG. 9 is an overall block diagram of a digital data over voice systemof the invention.

FIG. 10 is a block diagram of the timing recovery circuit of theinvention.

FIG. 11 is a detailed schematic of the coupling and filter circuits ofthe invention.

FIG. 12 is a partial schematic partial block diagram showing anembodiment in which a bridging phone is coupled into the system.

DESCRIPTION OF THE INVENTION I. THEORETICAL BACKGROUND OF THE INVENTION

Consider a data signal [a]given by an alphabet of finite values:

    [a]=[- - -, a.sub.-1, a.sub.o, a, , - - -]                 Equation 1

where;

a_(i) is an element of a finite number of discrete values; typically 0or 1. In baseband PAM transmission using a symbol p(t), the signal x(t)to be transmitted can be written as: ##EQU2## where; T is the reciprocalof the baud rate, or equivalently, the symbol period.

If p(T) is a Nyquist pulse, i.e., a pulse of the form shown in FIG. 1,then it should be apparent from Equation 2 that:,

    x(kT)=a.sub.k p(o)=Aa.sub.k                                Equation 3

Pulses of the type shown in FIG. 1 are called Nyquist pulses and havespecial spectral properties as discussed in "Digital Communication" byJohn A. Proakis, pages 338-341, McGraw Hill, 1983. Band-limited Nyquistpulses exist and can be used to signal through band-limited channels.Since copper pairs can only be equalized over a finite bandwidth, theyare good examples of band-limited channels. FIG. 2 shows the eyediagrams resulting from a "Raised Cosine" Nyquist pulse p(t) withFourier transform P(f) where: ##EQU3## and the data process is binary,i.e., a_(i) [0,1]. In FIG. 2, the current symbol is labelled CS, theprevious symbol PS, and the next symbol NS. It can be seen that the eyeis 100% open at the points t=kT, k {I}wherein I is the set of integerspositive, negative including zero. This is because the "raised cosine"pulses are Nyquist pulses and obey the property specified by Equation 3.

Approximations can be made to Nyquist Pulses in various ways. A commonmethod used in DDS is shown in FIG. 3, wherein a unit amplitude pulse Bof A volts amplitude is passed through a third order Butterworth filter10. The width of the pulse A is T/2 and the cutoff frequency fc of thefilter 10 is 0.65/T, which results in the formation of the pulse p_(F)(t). The little tail running past t=T would clearly interfere with thenext symbol and, therefore, the pulse is not completely Nyquist. On theother hand, the Inter Symbol Inference (ISI) caused by the little tailis relatively insignificant and the resulting eye is over 90% open. Sucha pulse is thus Almost Nyquist.

DDS transmits at 0.5 bits per hertz, or one bit every 2T seconds. Thus,DDS uses twice as much bandwidth as is necessary.

It can be shown (e.g., in "Signal Theory" by L.E. Franks, McGraw Hill,1969, page 13) that the power spectral density of the process describedin Equation 2 is of the form:

    S.sub.xx (f)=Y(f).P(f)                                     Equation 5

where P(f) is the Fourier transform of p(t) and Y(f) is some function offrequency.

For straight PAM, Y(f) consists of a continuous function plus impulsesat discrete frequencies. The continuous function is generally non-zeroat f=0. Since P(f), if it is a Nyquist pulse, is non-zero at zerofrequency, in general, S_(xx) (f), also contains D.C. These limitationsare overcome in DDS by AMI coding.

With the above theoretical background in mind, the theory behind thepresent invention, which involves the subject of sinusoidally CodedNyquist Pulses and sinusoidally Coded Almost Nyquist Pulses, will beexplained.

II. SINUSOIDALLY CODED PULSES

The following is a description of the coding method of the inventionwhereby a pulse shape can be modified to eliminate the energy in thevoice band. First, a brief digression into some mathematical constructsis necessary.

Let p(t) be a pulse function with Fourier Transform P(f). Define P_(m)(f), which is the Fourier transform of the sinusoidally encoded pulsep_(m) (t) as: ##EQU4## where T'=^(T).sbsp.4,m is an integer, and ∥P_(m)∥ is the normalizing function defined as: ##EQU5## Clearly, P_(m) (f)has unit energy as a result of the normalization. From Parsevals theoremp_(m) (t) is also a unit energy pulse. Further, it should be clear thatP_(m) (0)=0 and the first m-1 derivatives of P_(m) (f) with respect to fare also zero at f=0. Thus, by choosing m large enough, not only canD.C. be eliminated, but the lower frequency band can also be vacated.The extent to which the low frequency band is vacated as "m" isincreased is quantified below. To do this, a specific case for p(t) ischosen. The principles can then be applied to other cases.

Choose p(t) to be the least bandwidth, unit energy, Nyquist Pulse withFourier transform P(f) given by: ##EQU6## Thus, ##EQU7## and ##EQU8## Itshould be clear from the above that P_(m) (f) represents the spectralshaping accomplished by sinusoidal coding of the original pulse p(t);since the specific p(t) chosen in this case is spectrally flat. Also,since, ##EQU9## and applying this integral to Equation 10, it can beshown that: ##EQU10## Using the fact that:

    ∥P.sub.o ∥-1;                            Equation 13

one can interatively compute, ##EQU11## and so on.

FIG. 4 shows a power spectral density versus frequency/time plot of|P_(o) (f) (no coding case), |P₁ (f)|, |P₂ (f)|, |P₃ (f)| and |P₄ (f)|.By examination of these curves, it may be seen that |P_(m) (f)| tends toapproach an impulse function at ##EQU12## as m approaches infinity.

The coding gain CG, which is a measure of how much energy has beenvacated from the low frequency band by the coding function, is computedbelow; wherein, "m" represents the order of the coding function. The lowfrequency band may be defined as a fraction ρ of the total bandwidth1/T' of the signal P_(m) (f), i.e., the low frequency band is ##EQU13##where:

ρ is some real number less than one half. Then, the energy E_(m) (ρ) inthis low frequency band is given by: ##EQU14## Again using the standardintegral of Equation 10, it can be shown that: ##EQU15## In the presentexample, the pulse is assumed to be a unit energy flat spectra pulse,therefore:

    E.sub.o (ρ)=ρ                                      Equation 17

Using Equations 16 and 17, one can iteratively compute the energy, E_(m)(ρ) for any given fraction, ρ of the frequency band as "m" is increased.Since E_(o) (ρ) represents no coding, the coding gain (CG); which is ameasure of how much energy has been vacated from the low frequency bandas a result of coding, can be expressed as: ##EQU16## Using Equation 17,

    CG(ρ,m)=-10 log.sub.10 E.sub.m (ρ)+10 log.sub.10 ρ. Equation 19

measured in db. Table 1, below, summarizes CG(ρ,m) for ρ=1/4 and 1/14,respectively for m=0, 1, 2, 3 and 4.

                  TABLE 1                                                         ______________________________________                                        Fraction of Total  Fraction of Total                                          Bandwidth          Bandwidth                                                  ρ = 1/4        ρ = 1/14                                                    Energy     Coding Gain                                                                              Energy   Coding Gain                               m    E.sub.m (ρ)                                                                          CG(ρ,m)db                                                                            E.sub.m (ρ)                                                                        CG(ρ,m)db                             ______________________________________                                        0    0.25       0          7.143 × 10.sup.2                                                                 0                                         1    4.542 × 10.sup.-2                                                                  7.4        1.187 × l0.sup.-3                                                                17.8                                      2    1.417 × 10.sup.-2                                                                  12.5       3.538 × 10.sup.-5                                                                33.0                                      3    5.179 × 10.sup.-3                                                                  16.8       1.261 × 10.sup.-6                                                                47.5                                      4    2.045 × 10.sup.-3                                                                  20.9       5.562 × 10.sup.-8                                                                61.1                                      ______________________________________                                    

From Table 1, it may be seen that as ρ becomes smaller, the coding gainincreases more and more dramatically with increasing m.

In summary, by encoding the Fourier Transform of Nyquist or AlmostNyquist pulses with a sinusoidal function of the form Sin^(m) θ,spectral shaping of the pulses is achieved. This results in the removal,with increasing "m", of a greater and greater fraction of the energycontent from the low frequency base band to a higher frequency band.

III. Formation of Sinusoidally Coded Pulses

In accordance with the invention, a method will now be described wherebysinusoidally coded Nyquist or Almost Nyquist pulses can be formed fromthe original pulse p(t) by forming linear combinations of p(t) and itsdelayed and advanced versions p(t-kT), wherein k is an element of theset of integers I . This results in a simple circuit implementation of asinusoidal encoder, an embodiment of which will be described later.

Applying Euler's identity to Equation 6, it should be clear that:##EQU17## Now, using the Binomial Therorem, Equation 20 becomes:##EQU18## where ##EQU19## and i is an integer indicating the ith versionof the delayed pulse and ! is the factorial sign. Taking the FourierInverse of Equation 21 yields the desired result: ##EQU20## which is ofthe form expected above. Also note that Equation 22A can be written as:##EQU21## From Equation 22B, it should be clear that the pulse P_(m) (t)consists of delayed and advanced versions of p(t) corresponding to them+1 terms in Equation 22B. These versions are such that the i term isweighted by ##EQU22## As an example, it is shown below that for the casem=4, there are m+1=5 terms; the weighting on the 3^(rd) term (i=2 case)is therefore ##EQU23## is 6K, which checks with Equation 22₄ below.Expanding Equation 22A for m=0,1,2,3 and 4, one obtains: ##EQU24##

A voltage waveform of the type described by P₄ (t) in Equation 22 isdepicted in FIG. 5 wherein at t=-T' to -T'/2 the voltage amplitude A is1/4, at t=-T'/2 to 0, A is -1/2, at t=0 to +T'/2; A is +3/2, at t=+T'/2to T', A is -1/2 and at T=+T' to 3T'/2 A is +1/4.

The above theory is next applied to a specific example of creatingsinusoidally coded Nyquist Pulses. It will be demonstrated that bylooking in the time domain good (≧70%) eye opening can be obtained evenin the presence of sinusoidal encoding. Thus, baseband sinusoidallyencoded data can be mixed with baseband telephone voice with all theadvantages of high performance, high speed, low cost and low cross-talk.

Let p(t) be a double frequency Nyquist Pulse of the form used in DDS,i.e., ##EQU25## Recall that if the data signal for transmission (whichmay already be AMI encoded) was {- - - ,a₋₁, a_(o), a₊₁, - - - }then thePAM signal transmitted was: ##EQU26## and, hence, in view of Equation23; ##EQU27## Equation 24 is the reason why the eye was 100% open anddata could be recovered with no ISI.

Now consider the PAM process; ##EQU28## which, using Equation 21, can berewritten as: ##EQU29## Recalling that T'=T/4, one can again rewriteEquation 26 as: ##EQU30## where ##EQU31## Now, four cases can beseparated: ##EQU32## Expanding Equation 27 on case 1 and using Equation23, one writes: ##EQU33##

Similar operations on Equation 27 for case 2,3 and 4 using Equation 23can be used to show: ##EQU34##

Clearly, Equation 27 can be evaluated at different time points insteadof at ##EQU35## upon which Equations 28A, B, C and D were based. Forexample, it can be shown that: ##EQU36## and so on.

In conclusion, it has been shown that forming Nyquist or Almost Nyquistpulses into waveforms of the type described in Equation 22 is theequivalent of sinusoidally encoding Nyquist or Almost SinusoidallyNyquist pulses. Furthermore, sinusoidal coding results in knowninter-symbol interference, since all the coefficients in Equations 28A,B, C, D and 29 are known. The following example shows that knowledge ofthe ISI can result in very simple receiver structures not requiringadaptive techniques, such as Viterbi decoders, etc. to establishreliable communication.

IV. SINUSOIDAL ENCODER

A practical realization of sinusoidal encoding is shown in FIG. 6. Thefunction of the sinusoidal encoder 600 of FIG. 6 is to sinusoidallyencode the pulses P_(m) (t) of the input data stream as a function ofsin^(m) θ to redistribute the frequency of the energy in the data pulsesfrom the voice band to a higher frequency. As shown in Part III, thisencoding can be accomplished by forming appropriately weighted linearand delayed versions of the original pulse in accordance with the degreeof redistribution required. More specifically, Equation 22 shows thatp_(m) (t) may be formed of m+1 delayed and advanced versions of P(t).The delayed versions are the terms in Equation 22 corresponding to therange of i=0 to i=the integer less than or equal to m/2 and the advancedversions are the terms corresponding to i=the integer greater than orequal to m/2. The particular weighting of the versions is dependent on mand is shown in Table II below for i=0 to m and specific m's=1 to 4 andthe general case m=m.

                                      TABLE II                                    __________________________________________________________________________                                         General                                                                       Case                                     i m = 1   m = 2    m = 3    m = 4    m = m                                    __________________________________________________________________________       ##STR1##                                                                              ##STR2##                                                                               ##STR3##                                                                               ##STR4##                                                                               ##STR5##                                   ##STR6##                                                                              ##STR7##                                                                               ##STR8##                                                                               ##STR9##                                                                               ##STR10##                               2                                                                                        ##STR11##                                                                              ##STR12##                                                                              ##STR13##                                                                              ##STR14##                               3                                                                                                 ##STR15##                                                                              ##STR16##                                                                              ##STR17##                               4                                                                                                          ##STR18##                                                                              ##STR19##                               .                                    .                                        .                                    .                                        .                                    .                                        i                                                                                                                   ##STR20##                               .                                    .                                        .                                    .                                        .                                    .                                        m                                                                                                                   ##STR21##                               __________________________________________________________________________

ln the circuit of FIG. 6, shift register 20 provides the delayed andadvanced versions. Tapped weighting resistors R_(m) - R_(o), incombination with scaling factor resistors 28 and 30, perform theweighting function and summer 26, the summing function.

The pulsed data in the form of PAM raw data pulses or bits - - - a₋₁,a₀, a₁ - - - , to be encoded, is clocked into an n-stage CMOS shiftregister 20, the stages of which are coupled to respective weightingresistors R_(m) to R_(o), clocked with a clock of frequency 2/T, so thatevery bit is clocked-in twice. The output impedance of each cell orstage of the shift register has an impedance greater than or equal to 50ohms. The scale factor resistors (labelled 28 and 30) are chosen to makethe minimum resistor R_(min) Δ Minimum [R_(i), i=0, 1, - - - m] suchthat R_(min) is far, far greater than 50 ohms.

The pulse response of such a circuit is P_(m) (t) and since the circuitis linear, the output s_(m) (t) is: ##EQU37## as desired.

In the circuit of FIG. 6, each original pulse p(t) in the data stream orsequence 22 is combined with its delayed or advanced versions p(t-kT),where k is an element of the set of integers I. The delayed and advancedversions are combined with the original pulse to produce a sinusoidallyencoded pulse waveform of the form sin^(m) θ; wherein m is an integergreater than or equal to 1. The resultant encoded pulse is of the formshown in FIG. 5 for m=4. This is accomplished by coupling the datastream 22 to the input of the above-referenced shift register 20,successive stages 24 of which are coupled to output voltage dividernetworks.

The voltage divider networks comprise weighting resistors R_(m),R_(m-1), R_(m-2), R_(m-3) - - - R_(i) - - - R₁, R_(o) wherein m=theorder or level of encoding and is equal or greater than 1 and i is aninteger corresponding to the ith delayed pulse. The weighting resistorsare alternately coupled to a precise positive potential (+Volt Rail) anda precise negative potential (-Volt Rail). The voltage divider networkfurther includes resistor 28 equal to R/ ∥"Pm"∥ tied between the plusRAIL and ground; the junction of which is coupled to the positive inputterminal of difference amplifier 26; and also includes resistor 30 tiedbetween the negative RAIL and the output of difference amplifier 26. Thenegative RAIL is also coupled to the negative input terminal of summingamplifier 26 and the positive RAIL is coupled to the positive inputterminal.

The generalized magnitude of the weighting resistors is shown in TableIII, below, and for the specific cases of m=1 through 4, in TABLE II,previously noted.

                  TABLE III                                                       ______________________________________                                                           i                                                          ______________________________________                                         ##STR22##           m                                                         ##STR23##           m-1                                                       ##STR24##           m-2                                                       ##STR25##           m-3                                                      .                    .                                                        .                    .                                                        .                    .                                                         ##STR26##           i-1                                                       ##STR27##           i                                                        .                                                                             .                                                                              ##STR28##           1                                                         ##STR29##           0                                                        ______________________________________                                    

The output of difference amplifier 26 is therefore equivalent to thesinusoidal encoding of the Fourier transform of the input data sequencepulses wherein each pulse p(t) has been linearly combined with itsdelayed and its advanced versions such versions being contained in thesuccessive stages 24 of the shift register 20. The form in which theyare combined is a function of the weighting resistors. In the case ofm=4, they are combined as a function of (1-cosθ)² to produce pulsewaveforms of the type previously shown in FIG. 5. In accordance with thediscussion in part III, it has been shown that such coding removes theenergy content of the input pulses from baseband to a higher frequencyas "m" is increased. These coded pulses are then coupled through pulseresponse circuit, such as a third order Butterworth filter 32, fortransmission over cable pairs to a suitable receiver described below.

FIG. 7 is an alternate embodiment, in which like items carrycorresponding numeral references with a prime suffix. The FIG. 7embodiment shows the case of concatenated encoding, i.e., a precodingfunction, such as AMI, followed by sinusoidal encoding. Such precodingmay produce other desirable spectral characteristics, such as a ZERO, atmidband, achieved by partial response coding. In the case of AMIpre-coding, note that a ZERO is produced at D.C. and the spectralcharacteristics are shaped by a sine function; thereby helping tofurther eliminate energy from the low frequency band to be used forvoice (POTS) frequency signalling. The pre-coder 40 breaks the AMI/PAMdata signal 22' into two pulse rails, a plus pulse rail 42 and minuspulse rail 44, which are then separately encoded, as in FIG. 6, by theirrespective individual shift registers 46 and 48 and summed inoperational amplifier 26' and coupled to filter 32'.

V. SINUSOIDAL DECODING

A simple sinusoidal decoder will now be described for the specific caseof m=4 sinusoidally encoded pulse communication.

Consider the case m=4, i.e, l=1 for which Equation 29 can be written as:##EQU38## wherein a_(o) is the current bit and a₋₁ is the previous bit.Note that the next bit a₊₁ does not appear in Equation 31.

Now, consider the case where the sequence {- - - , a₋₁, a_(o), a₁, - - -} has already been AMI encoded. For this case, Equation 30 can only havethe following values: ##EQU39##

A receiver which satisfies the conditions of Equation 31 is shown inFIG. 8. A sinusoidally encoded AMI/PAM received signal s₄ (t) of theform shown in FIG. 5, is sampled by window comparator 50 to determine ifit lies between + or -2KA; wherein K is a scaling factor ##EQU40## i.e.,for m=4,∥P_(m) ∥=0.523 (Equation 14) and K=0.239A; and A=peak amplitudeof the symbol being used. Since A is known in advance, this informationcan be used in the receiver to create maximum power conditions. If theanswer from the window comparator 50 is "Yes", a one pulse istransmitted from comparator 50 to logic circuit 52 to provide one of twoinputs to EX-OR Gate 54. The other input terminal of EX-OR Gate 54 iscoupled to the output of Delay Flip-Flop 56 which is clocked at timeintervals corresponding to t=-T/2+kT. Flip-Flop 56 produces an outputwhich is a function of the previous decision (a[(k-1)T], i.e., theD-Input to Flip-Flop 56 that occurred one clock pulse earlier. Theoutput of EX-OR 54 is coupled back to the D-terminal of Flip-Flop 56 toprovide the previous decision. Thus, the EX-OR 54 output is a(kT) andcoincides with a clock pulse at t=T/2+kt. From Equation 31 and FIG. 6,it should now be clear that:

    s.sub.4 (-T/2+kt)=a[(k-1)T]⊕a(kt)                      Equation 33

where: Δ means an Exclusive/Or function. Inverting Equation 33, it canbe seen that:

    a(kt)=a[(k-1)T]⊕s.sub.4 (-T/2+kT)                      Equation 34

Thus, once the correct start-up condition, a_(o), has been set up, thereceiver will continue to generate the correct sequence {a_(k), k=0,1, - - - }. As can be recognized, this receiver can cause infinite errorpropagation if a single error is made. This can be resolved by locatingthe Exclusive/OR function at the transmitter prior to AMI encoding.Alternatively, in a system where a_(k) can be forced to a knowncondition periodically, for example, in a time-coded multiplex (TCM)system, prior Exclusive/OR encoding at the transmitter is not necessarysince the errors will propagate only until the next forcing instant.Thus, in a TCM system, errors can be made to propagate only for a frame.

VI. DDOV SYSTEM

FIG. 9 illustrates a Digital Data Over Voice (DDOV) system of theinvention in block diagram form. In this embodiment, voice frequencysignals from a voice frequency communication system, such as telephoneset 700, at a frequency below 4 kHz originating at Customer PremisesEquipment (CPE) are transmitted from coupling circuit C1 along withsinusoidally encoded bursts of Time Compressed Multiplexed (TCM) datasignals. TCM logic circuit 100 at the CPE, using well-known techniques,accepts serial Transmit Data and Transmit Clock pulses at base-bandfrequency, performs the TCM function and transmits clock pulses and datapulses 108' in bursts of short duration. Correspondingly, TCM logiccircuit 200 performs a similar function at the Central Office Equipment(COE); and data and timing signals 208' are transmitted in the oppositedirection from coupling circuit C2 over wire pair 108 in the oppositedirection toward C1.

Typically, the CPE station is slaved to the COE station so that themaster station (COE) transmits first in one burst 208' during one timeslot in one direction, then the slave transmits in one burst 108' in alater time slot. A time slot gap in transmission of time durationgreater than the round trip delay of the longest anticipated cable isprovided between transmissions to avoid collision. One burst from themaster, plus a corresponding burst from the slave, together with thegap, constitute a "frame". Then the process commences again withtransmission of another frame. This process is called "ping-pong"communication.

In a specific embodiment, the data transmission format for each burst108' or 208' may comprise 56 bits of data preceded by a two bit timingheader of successive ONE'S and followed by a tail of two bits for atotal of 60 bits per burst in each direction. A typical time gap betweenbursts is 20 bits in length, resulting in a total frame of 140 bits;which at 56 Khz, is 2.5 m sec. duration and a frame rate of 400 Hz.

The tail in the format consists of an ODD parity bit followed by a ONE.With ODD parity, the total number of ONES in the burst, up to andincluding the parity bit itself, is forced by the TCM TransmissionLogic, to be an odd number by appropriately making the parity bit a ZEROor a ONE. Therefore, since the last bit in the tail is a ONE, everyburst has an even number of ONE'S. Hence, every burst begins by the samedeviation transition, despite AMI encoding. The result is ZERO D.C. perburst, which results in no bipolar violations from frame-to-frame.

A Sinusoidal Encoder 102 or 202, at the CPE or COE, respectively, of thetype shown and described in FIG. 6, encodes the clock and data pulses toempty the energy content of the clock and data pulses from the voicefrequency band to a higher frequency band.

The encoded clock and data pulses are coupled through a High Pass Filter204 or 104 at the CPE or COE, respectively, having cut-off frequency at8 kHz and pass to coupling circuit C1 or C2, respectively, for couplingto a single pair of standard telephone wires forming a local loop cable108. During transmission, analog switches SW1 and SW2 are opened toprevent transmitted data from being coupled into the respectivereceivers 110 or 210.

Simultaneously, the low frequency voice signal (VOICE) is coupled in thenormal manner through low pass filters 106 or 206 to coupling circuitsC1 or C2 and across the same cable 108; which cable forms the standardtip and ring conductors for voice telephone communication. Filters 106or 206 prevent the data signals from interfering with the voice signaland high pass filters 104 or 204 perform a similar function preventingany residual energy of the data signal remaining in the 0-4 kHz bandfrom being passed out to the local loop and interfering with the voicesignals.

At the COE, the sinusoidally encoded data and clock pulses are coupledfrom HPF filter 204 to a cable equalizer 220, wherein the signals arecompensated for cable losses associated with the cable 108, as describedin the aforementioned patent application. After compensation, thesignals are coupled to data receiver 210, which is of the type describedin FIG. 8 to recover the original clock and pulse data generated fromTCM 100. The recovered clock and data signals are demultiplexed andreturned to their pre-burst form in (Master) TCM LOGIC Circuit 200.

Similarly, voice signals in the opposite direction are coupled throughC1 to LPF 106 to a voice frequency communication system, in this case, avoice switch (not shown), while burst data is coupled from C1 throughHPF 104 through closed switch SW1 to equalizer 120 and decoded in datareceiver 110.

The header bits, parity bits, and the trailing 1 bit (called "tail") areused in the timing recovery circuit 109 to recover the master dataclock, which is then used to synchronize the TCM logic circuits in awell-known manner. Referring to FIG. 10, the timing recovery circuit 109comprises a slicer circuit 900, which determines when the equalizedincoming signals exceeds 2KA and is therefore a data signal. When theincoming signal exceeds the threshold level of 2KA, a one shotmultivibrator 902, having a time constant greater than the burst period,is triggered to provide the leading edge and lagging edge of the burstperiod for gating a phase lock loop circuit 104, which regenerates theclock pulses for synchronization. The format of the data signals enablesthe timing recovery circuit to provide the burst rate from which the PLLin the slave station derives the received clock. Other standard timingrecovery algorithms can also be employed.

VII. COUPLING CIRCUITS

Additional details of the coupling circuits C1 and C2, which perform theline coupling, filtering, return loss optimization, and bridgingfunctions of the invention will now be described in connection with FIG.11.

In the circuit of FIG. 11, one coupling circuit C2 and low pass circuit206 are shown in detail. It should be noted that at the customerspremises equipment (CPE) identical circuits C1 (FIG. 9) and 106 areprovided. The circuit C2 couples incoming and outgoing voice and datasignals across the tip (108T) and ring (108R) lines of the local loopcable 108 with appropriate impedance matching at the frequency band ofeach.

Positive and Negative encoded data signals for transmission are coupledfrom amplifiers 702 and 704, respectively, to impedance matchingresistors R1 and R2 coupled across the primary windings of pulsetransformer T1 of coupling circuit C2. Amplifier 702 amplifies theuninverted positive transmit encoded pulses generated by the encoder ofFIG. 6 or FIG. 7. For balanced line transmission, the positive pulsesare inverted and amplified in amplifier 704.

Resistors R1 and R2 have an impedance of 67.5 ohms, each, for a total of135 ohms, satisfying the requirement that the outgoing data pulses see acharacteristic impedance of 135 ohms at the data frequency. Ideally, thevoice frequency band should be matched to a characteristic impedance of900 and 2.15 microfarads at the voice frequencies.

Capacitor C12, having a capacity of 0.039 microfarads, is coupledbetween the secondary windings of balanced transformer T1. At voicefrequencies, C1 is part of a low pass voice frequency filter; whereas atdata frequencies, C1 is an effective short circuit, so that T1 at datafrequencies is essentially a 1:1 transformer. Furthermore, at the datafrequencies, the inductors L4A and L4B, in series with the secondarywindings of T1, represent a high impedance, thereby buffering the datasignals from the telephone load seen through the low pass filter stages.

In order to avoid the passage of ringing current to the telephonereceiver 700, the coupling circuit and filter must be carefullystructured to avoid unbalancing the signal. Thus, the data signal iscoupled using a balanced transformer T1 to keep earth/ground parasiticsbalanced and is then coupled to a difference amplifier 706 to a balancedreceiver and high pass filter 204 (FIG. 9).

The ringing signal on the local loop has a longitudinal signal componentand a metallic signal component. Inductors L5A and L5B, together withC9, C10 and C11 in parallel across the line, form a second order lowpass filter, which is transparent to the metallic component of ringingcurrent and suppresses the longitudinal component of ringing circuit.Preferred values for each component are as follows:

L5A - 25 millihenry

L5B - 25 millihenry

C9 - 0.051 microfarad

C10 - 0.056 microfarad

C11 - 0.022 microfarad

The metallic filter is comprised of four parallel coupled capacitors andinductors C5, L1A; C7, L2A; C6, L1B; and C15, L2B having the followingpreferable values:

C5 - 0.022 microfarad

C7 - 0.033 microfarad

C6 - 0.022 microfarad

C15 - 0.033 microfarad

L1A - 8.5 millihenry

L2A - 8.8 millihenry

L1B - 8.5 millihenry

L2B - 8.8 millihenry

Suppressing the longitudinal ringing component extends the useful rangeof the system above 12-13 kilofeet of 26 gauge copper loops to at least18 kilofeet of 26 gauge copper wire.

It should be noted that the impedance matching structure, abovedescribed, results in an optimum average return loss over the lowfrequency band of the voice signals 200 Hz to 1.0 kHz, the mid-frequencyband of 500 Hz to 2500 Hz and the high frequency band of 2 kHz to 3.3kHz.

The return loss, which is a measure of the imperfection in the impedancetermination, is a function of frequency. At the low and highfrequencies, singing return loss must not be greater than 10 db at themid-frequencies and echo return loss must not be greater than 18 db.Computer optimization, together with the constraint of practicalcomponent values, have gone into generating the values of the componentsdescribed above. These component values result in singing return lossesat low and high frequencies, to exceed 20 db, while echo return lossexceeds 26 db.

When it is desired to bridge a second phone 701 (See FIG. 12) across thetelephone lines (108T and 108R), isolation is required to preserve voicefrequency impedance matching when the primary phone 700 is OFF-Hook,thereby changing the load on the line. Also, the second phone 701 has tobe equipped with a low pass filter (LPF) 206'.

For isolation purposes, each phone is provided with a saturable ferritereactor with very high inductance. The reactor is formed of respectivetightly coupled inductors L3A and L3B and L3A' and L3B' coupled togetherby ferrite cores FC1 and FC1' forming a composite coupled inductanceexceeding 2H. When either phone 701 or 700 is OFF-Hook, DC current flowsthrough its reactor windings L3A' and L3B' or L3A, L3B, respectively.Its reactor saturates at 1 or 2 milliamps of current, effectivelyeliminating itself from the circuit. The reactor in front of the ON-Hookphone is not saturated. When the OFF-Hook phone is placed ON-Hook, DCcurrent flow stops and the inductors L4A and L4B, or L4A', L4B', whichare fairly large, i.e., much greater than 0.5 Henry, do not effect thetuning of the low pass filter at the voice frequency.

Equivalents

This completes the description of the preferred embodiments of theinvention. Those skilled in the art may recognize many variationsthereof. The invention should not be limited, except as required by thescope of the following claims and equivalents thereof.

What is claimed is:
 1. Apparatus for encoding data signals fortransmission at a predetermined rate over telephone lines in thepresence of voice band telephone signals such that the frequencyspectrum of the data signals is substantially removed from the voiceband and is shifted to a higher frequency spectrum comprising:(a)generator means for generating pulsed data signals in bit intervals ofduration T; (b) coding means for forming linear combinations of saidpulsed data signals with delayed and advanced version thereof tosinusoidally encode the pulses in the frequency domain by convertingeach data bit into a corresponding data symbol having a duration 5T/2,the data symbol being composed of five pulses spaced T/2 apart havingalternately opposite polarities and equal durations of not more thanT/2, the amplitudes of a first pair of pulses on either side of acentral pulse being equal and being "a" times an amplitude of thecentral pulse, where "a" is a factor ≦1, and the amplitudes of a secondpair of pulses on either side of the first pair of pulses being equaland being "b" times the amplitude of the central pulse where "b" is afactor ≦1.
 2. The apparatus of claim 1 wherein the data signals are inthe form of substantially Nyquist pulses.
 3. A transmission systemcomprising:(a) a transmitter for transmitting binary data signals via atransmission means, the transmitter comprising:(i) a data source forgenerating data bits in synchronous consecutive bit intervals having aduration T; and (ii) a code converter for converting each data bit intoa corresponding data symbol; and (b) a receiver comprising:(i) areceiver filter; and (ii) a data detector for detecting the transmitteddata symbols: wherein the improvement comprises that: (c) thecode-converter is arranged to convert each data it into a correspondingdata symbol having a duration 5T/2, the data symbol being composed offive pulses of T/2 duration having alternately opposed polarities, theamplitudes of a first pair of pulses on either side of a central pulsebeing equal and being "a" times an amplitude of the central pulse, where"a" is a factor≦1, and amplitudes of a second pair of pulses on eitherside of the first pair of pulses being equal and being "b" times theamplitude of the central pulse where "b" is a factor≦1.
 4. Atransmission system comprising:(a) a transmitter for transmitting binarydata signals via a high-pass transmission means, the transmittercomprising:(i) a data source for generating data bits in synchronousconsecutive bit intervals having a duration T; and (ii) a code converterfor converting each data bit into a corresponding data symbol ofduration 5T/2, the data symbol being composed of first, second, third,fourth, and fifth pulses, each pulse being of equal pulse duration, saidpulse duration being not more than T/2, said pulses being equallyspaced, and said pulses being of alternately opposite polarities, saidthird pulse being a central pulse having an amplitude, said second andfourth pulses forming a first pair of pulses on either side of thecentral pulse, said first pair of pulses having equal amplitudes whichare a factor of "a" times the amplitude of the central pulse, where "a"is a number less than or equal to one, the first and fifth pulsesforming a second pair of pulses on either side of the first pair ofpulses, the second pair of pulses having equal amplitudes which are afactor of "b" times the amplitude of the central pulse, where "b" is anumber less than or equal to one, whereby transmitted data symbolsoverlap to form a multi-level transmission signal; and (b) a receivercomprising:(i) a receive filter; and (ii) a data detector for detectingthe transmitted data symbols.
 5. A transmission system as claimed inclaim 3 wherein "a " is 0.6667 and "b" is 0.1667.
 6. A code convertercomprising:(a) an input for receiving data bits in synchronousconsecutive bit intervals of duration T; (b) means for converting eachdata bit into a corresponding data symbol of duration 5T/2, the datasymbol being composed of first, second, third, fourth, and fifth pulses,each pulse being of equal pulse duration, said pulse duration being notmore than T/2, said pulses being equally spaced, and of alternatelyopposite polarities, said third pulse being a central pulse having anamplitude, said second and fourth pulses forming a first pair of pulseson either side of the central pulse, said first pair of pulses havingequal amplitudes which are a factor of "a" times the amplitude of thecentral pulse, where "a" is a number less than or equal to one, thefirst and fifth pulses forming a second pair of pulses on either side ofthe first pair of pulses, the second pair of pulses having equalamplitudes which are a factor of "b" times the amplitude of the centralpulse, where "b" is a number less than or equal to one; (c) an output atwhich said data symbol is provided.
 7. The converter of claim 6 wherein"a" is 0.6667 and "b" is 0.1667.
 8. A receiver for receiving datasymbols representing binary data signals, including data bits havingconsecutive bit intervals of duration T, the receiver comprising:(a) aninput for receiving data symbols which are composed of first, second,third, fourth, and fifth pulses, each pulse being of equal pulseduration, said pulse duration being not more than T/2, said pulses beingequally spaced, so that sequential pulses are 5T/2 apart, and saidpulses being of alternately opposite polarities, said third pulse beinga central pulse having an amplitude, said second and fourth pulsesforming a first pair of pulses on either side of the central pulse, saidfirst pair of pulses having equal amplitudes which are a factor of "a"times the amplitude of the central pulse, where "a" is a number lessthan or equal to one, the first and fifth pulses forming a second pairof pulses on either side of the first pair of pulses, the second pair ofpulses having equal amplitudes which are a factor of "b" times theamplitude of the central pulse, where "b" is a number less than or equalto one; (b) a receive filter coupled to the input; and (c) a datadetector coupled to an output of the receive filter for detecting thedata symbols.
 9. A transmitter for transmitting binary data signals, thetransmitter comprising:(a) a data source for generating data bits insynchronous consecutive bit intervals having a duration T; (b) a codeconverter for converting each data bit into a corresponding data symbolof duration 5T/2, the data symbol being composed of first, second,third, fourth, and fifth pulses, each pulse being of equal pulseduration, said pulse duration being not more than T/2, said pulses beingequally spaced, so that sequential pulses are 5T/2 apart, and saidpulses being of alternately opposite polarities, said third pulse beinga central pulse having an amplitude, said second and fourth pulsesforming a first pair of pulses on either side of the central pulse, saidfirst pair of pulses having equal amplitudes which are a factor of "a"times the amplitude of the central pulse, where "a" is a number lessthan or equal to one, the first and fifth pulses forming a second pairof pulses on either side of the first pair of pulses, the second pair ofpulses having equal amplitudes which are a factor of "b" times theamplitude of the central pulse, where "b" is a number less than or equalto one.
 10. The transmitter of claim 9 wherein "a" is 0.6667 and "b" is0.1667.